Synchronous motor driving system

ABSTRACT

Axial error calculation unit is provided for estimating an axial error Δθ between a d-q axis and a dc-qc axis by using Ld, Lq, Ke, Id*, Iq*, Idc and Iqc in a range of all rotational speeds except zero of a rotational speed command of a synchronous motor, Ld denoting an inductance on a magnetic pole axis d of the synchronous motor, Lq an inductance on a q axis orthogonal to the magnetic pole axis d, Ke a generated power constant of the motor, Id* a current command of the d axis, Iq* a current command on a q axis, Idc a detected current value on an assumed dc axis on control, and Iqc a detected current value on an assumed qc axis orthogonal to the assumed dc axis. Irrespective of presence of saliency, position sensorless control can be achieved in a wide range a low to high speed zone.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a synchronous motor driving system, andmore particularly, it relates to a control method for achieving a highlyprecise and high-performance synchronous motor driving system withoutusing any sensors for detecting a rotational speed of a synchronousmotor and a position of a magnetic pole.

2. Description of the Related Art

Many developments have been made for a method for controlling asynchronous motor without detecting the rotational speed of thesynchronous motor and the position of a magnetic pole. Such controlmethods are usually classified into two types.

A first type is a control method based on speed/position sensor vectorcontrol of the synchronous motor. Instead of using the speed/positionsensor, a magnetic pole position estimating instrument, and a speedestimating instrument are used. For example, a method is known, which isdescribed in a document 1: “No. SPC-00-67: A New Position SensorlessControl of IPM Synchronous Motor using Direct Position ErrorEstimation”, by inventors Kiyoshi Sakamoto, Yoshitaka Iwazi, andTsunehiro Endo, in “IEEJ Semiconductor Power Conversion/IndustrialElectric Power Application Joint Research Material” (November, 2000).This method is known as a vector control sensorless system.

A second type is a control method called a V/F control system, whichcontrols the synchronous motor by an open loop.

In the case of the vector control sensorless system, except fornon-presence of a position/speed sensor, a configuration itself of thecontrol system is similar to that of a vector control system equippedwith a sensor. Accordingly, a high-performance synchronous motor drivingsystem can be achieved.

FIG. 15 shows a relation between a d-q coordinate axis and an assumedaxis dc-qc by a vector with a magnetic pole axis of a synchronous motorset as a reference axis. For vector control, as shown in FIG. 15, amagnetic pole axis of the synchronous motor is set as a d axis, an axisorthogonal to the same as a q axis. Then, by properly controlling avoltage and a current applied to the synchronous motor on each axis,high-performance making utmost use of synchronous motor performance isachieved. According to this vector control sensorless system, torque canbe made linear, and efficiency can be maximized.

In the case of the vector control sensorless system, a dc-qc axis is setby assuming a d-q axis on control, deviation (axial error) Δθ from areal d-q axis is estimated, and a dc-qc axial phase is adjusted toreduce the deviation to zero. Thus, in the case of the vector controlsensorless system, a method of estimating an axial error Δθ is a mostimportant factor for deciding control performance.

In well-known examples, several estimation methods of axial errors Δθhave been presented according to the rotational speed zones of thesynchronous motor. In practice, all the speed zones are covered by usingthese control methods in association.

On the other hand, in the case of the V/F control system, no speed orcurrent automatic adjustment units are provided, and a voltage to beapplied to the synchronous motor is decided. As its conventionalexample, a control method is described in JP-A-2000-236694. In the caseof the V/F control, different form the case of the vector sensorlesssystem, a magnetic pole axis is not estimated. Thus, a configuration ofa control system is greatly simplified. However, if a load is suddenlychanged during driving, transient vibration may occur. In order tosuppress such transient vibration, JP-A-2000-236694 presents a controlsystem for correcting a speed based on a current detected value.

In the case of the vector control sensorless system, a sensorless systemmust be switched to another according to a driving speed of thesynchronous motor. The method described in the document 1 estimates anaxial error Δθ based on a speed electromotive voltage of the synchronousmotor in principle, which can be achieved only in a middle/high speedzone. A similar problem is inherent in the method described inJP-A-8-308286.

On the other hand, as a sensorless system of a low speed, many methodshave been presented, which uses a inductance difference (saliency) ofthe synchronous motor. For example, as described in JP-A-7-245981, thereis a method for superposes a higher harmonic wave on a voltage command,and calculates an axial error based on a higher harmonic currentcomponent thus generated.

In this method, however, since it is necessary to superpose the higherharmonic wave, a pulsation component is generated in a synchronous motorcurrent, causing a considerable reduction in efficiency of thesynchronous motor. In addition, because of the superposed wave,electromagnetic noise is increased. Especially, to detect an axial errorwith high sensitivity, the amount of superposed waves to be injectedmust be increased, and thus it is difficult to solve the above-describedproblems, and achieve high control performance at the same time.

In addition, since the saliency of the synchronous motor is used, thesystem cannot be applied to a synchronous motor of a non-saliency type.Further, when the low-speed system, and the middle/high speed system areused in association, the systems must be switched according to a speed,and thus shocks occur following the switching.

On the other hand, in the case of the V/F control, the synchronous motorcan be driven from a low to high speed zone by a configuration of asingle control system.

However, in the V/F control, since the d-q axis in the synchronous motoris not basically coincident with the dc-qc axis on control, it isdifficult to achieve high-performance control. For example, it isdifficult to achieve high-speed response to a change in a rotationalspeed command, linear control of torque, maximum efficiency control andthe like. Accordingly, there is a possibility that external disturbancessuch as fluctuation in load torque may cause inconveniences such asvibration or excessive current.

SUMMARY OF THE INVENTION

An object of the present invention is to provide a synchronous motordriving system equipped with means for stabilizing a control systemwithout superposing any higher harmonic waves, controlling a low-speedzone to a high-speed zone by a continuous method, and achieving a vectorcontrol sensorless system.

The foregoing object of the present invention can be achieved byproviding means for calculating an error Δθ as deviation between amagnetic pole axis in a synchronous motor and a magnetic pole axis oncontrol, as a function of a driving current command, a detected currentvalue, an inductance constant, and generated power constant of thesynchronous motor, calculating an axial error by applying the axialerror calculating means to all speed zones of the synchronous motorexcept zero, and correcting the magnetic pole axis on the control basedon the axial error.

That is, in order to achieve object, in accordance with the presentinvention, there is provided a synchronous motor driving system whichcomprises a synchronous motor, an inverter for driving the synchronousmotor, a rotational speed command generator for supplying a rotationalspeed command to the synchronous motor, and a control unit forcalculating a voltage applied to the synchronous motor, said synchronousmotor driving system comprising axial error calculation means forestimating an axial error Δθ between a d-q axis and a dc-qc axis byusing Ld, Lq, Ke, Id*, Iq*, Idc and Iqc in a range of all rotationalspeeds except zero of the rotational speed command of the synchronousmotor wherein Ld is an inductance on a magnetic pole axis d, Lq is aninductance on a q axis orthogonal to the magnetic pole axis d, Ke is agenerated power constant of the motor, Id* is a current command of the daxis, Iq* is a current command on the q axis, Idc is a detected currentvalue on an assumed dc axis on control, and Iqc is a detected currentvalue on an assumed qc axis orthogonal to the assumed dc axis; and meansfor adjusting the dc-qc axis to the d-q axis based on the calculatedvalue of the axial error Δθ.

Furthermore, in order to achieve the above object, in accordance withthe present invention, there is provided a synchronous motor drivingsystem which comprises a synchronous motor, an inverter for driving thesynchronous motor, a rotational speed command generator for supplying arotational speed command to the synchronous motor, and a control unitfor calculating a voltage applied to the synchronous motor, saidsynchronous motor driving system comprising axial error calculationmeans for estimating an axial error Δθ as a function of an inductance Land a generated power constant Ke among the resistance R, the inductanceL and a generated power constant Ke as synchronous motor constants ofthe synchronous motor in a range of all rotational speeds except zero ofa rotational speed command of the synchronous motor; and means foradjusting the dc-qc axis to the d-q axis based on the calculated valueof the axial error Δθ, wherein Ld is an inductance on a magnetic poleaxis d, Lq is an inductance on a q axis orthogonal to the magnetic poleaxis d, Ke is a generated power constant of the motor, Id* is a currentcommand of the d axis, Iq* is a current command on the q axis Iq*, Idcis a detected current value on an assumed dc axis on control, and Iqc isa detected current value on an assumed qc axis orthogonal to the assumeddc axis.

The axial error calculation means can be adapted to calculate an axialerror Δθ by using current commands Id* and Iq* on the d-q axis, anddetected current values Idc and Iqc on the dc-qc axis, wherein Ld is aninductance on a magnetic pole axis d, Lq is an inductance on a q axisorthogonal to the magnetic pole axis d, Ke is a generated power constantof the motor, Id* is a current command of the d axis, Iq* is a currentcommand on the q axis, Idc is a detected current value on an assumed dcaxis on control, and Iqc is a detected current value on an assumed qcaxis orthogonal to the assumed dc axis.

In this case, instead of the detected current value Idc on the dc axis,a current command Id* can be used.

In addition, the synchronous motor driving system may be provided withmeans for detecting a DC current on a power source side of the inverter,and synchronous motor current estimating means for estimating an ACcurrent of the synchronous motor based on the detected DC current and adriving pulse signal for driving the inverter, and the axial error Δθmay be calculated using the estimated current as a detected currentvalue.

The synchronous motor driving system can be provided with means fordetecting a DC current on a power source side of the inverter, and Iqcestimating means for estimating a current value on the qc axis of thesynchronous motor based on the detected DC current, and a detected valueor a set value of a DC voltage of the inverter, and the axial error Δθcan be calculated using the estimated current as a detected currentvalue.

In any of the above-described synchronous motor driving systems, in thecalculation of the axial error Δθ, a correction term may be provided tomake correction in accordance with a rotational speed command of thesynchronous motor, and the correction term may be set as a function ofweight which increases as the rotational speed command approaches zero.

The current command Iq* on the q axis can be made based on the detectedcurrent value or the estimated value on the qc axis.

The synchronous motor driving system can be provided with means forestimating a speed deviation between the rotational speed command and areal rotational speed based on the calculated value of the axial errorΔθ, and the q axis current command Iq* of the synchronous motor can bemade based on the estimated value of the speed deviation.

The synchronous motor driving system may be provided with means forestimating a rotational speed of the synchronous motor based on thecalculated value of the axial error Δθ, and the q axis current commandIq* of the synchronous motor may be made based on a deviation betweenthe estimated value and the rotational speed command.

In the above-described synchronous motor driving systems of the threetypes, the current command Id* of the d axis can be made based on thecurrent command Iq* of the q axis.

In the present invention, the synchronous motor may be a salient type ora non-salient type.

Other objects, features and advantages of the invention will becomeapparent from the following description of the embodiments of theinvention taken in conjunction with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram showing a configuration of a synchronous motordriving system according to a first embodiment of the present invention.

FIG. 2 is a perspective view showing a schematic structure of thesynchronous motor driving system of the first embodiment.

FIG. 3 is a block diagram showing an internal configuration of a voltagecommand calculator 12 in the synchronous motor driving system of thefirst embodiment of the present invention.

FIG. 4 is a block diagram showing an internal configuration of an axialerror calculator 14 in the synchronous motor driving system of the firstembodiment of the present invention.

FIG. 5 is a block diagram showing a configuration of a synchronous motordriving system according to a second embodiment of the presentinvention.

FIG. 6 is explanatory view showing an operation of a synchronous motorcurrent estimator 18 in the synchronous motor driving system of thesecond embodiment of the present invention.

FIG. 7 is a block diagram showing a configuration of a synchronous motordriving system according to a third embodiment of the present invention.

FIG. 8 is a block diagram showing an internal configuration of an axialerror calculator 14D in a synchronous motor driving system according toa fourth embodiment of the present invention.

FIG. 9 is an explanatory view showing an operation of a functiongenerator in the synchronous motor driving system of the fourthembodiment of the present invention.

FIG. 10 is a block diagram showing an internal configuration of acontrol unit 2E in a synchronous motor driving system according to afifth embodiment of the present invention.

FIG. 11 is a block diagram showing an internal configuration of acontrol unit 2F when the synchronous motor driving system of the fifthembodiment of the present invention is applied to the third embodimentshown in FIG. 7.

FIG. 12 is a block diagram showing an internal configuration of acontrol unit 2G in a synchronous motor driving system according to asixth embodiment of the present invention.

FIG. 13 is a block diagram showing an internal configuration of acontrol unit 2H in a synchronous motor driving system according to aseventh embodiment of the present invention.

FIG. 14 is a block diagrams showing an internal configuration of acontrol unit 2J in a synchronous motor driving system according to aneight embodiment of the present invention.

FIG. 15 is a view showing a relation between a d-q coordinate axis witha magnetic pope axis of a synchronous motor set as a reference, and anassumed dc-qc axis on control by a vector.

DESCRIPTION OF THE EMBODIMENTS

Next, description will be made of the preferred embodiment of thesynchronous motor driving system of the present invention with referenceto FIGS. 1 to 15.

First Embodiment

FIG. 1 is a block diagram showing a configuration of a synchronous motordriving system according to a first embodiment of the present invention.The synchronous motor driving system of the first embodiment comprisesthe following elements: a rotational speed command generator 1 forsupplying a rotational speed command ωr* to the synchronous motor; acontrol unit 2 for calculating amplitude, a frequency and a phase of avoltage applied to the synchronous motor; a pulse width modulation (PWM)generation 3 for generating a pulse for driving an inverter 4 based on avoltage command V1*; the inverter 4 for driving the synchronous motor:the synchronous motor 5 to be controlled; a current detector 6 fordetecting a current of the synchronous motor 5; a conversion gain 7 forconverting the rotational speed command ωr* into an electrical anglefrequency command ω1* of the synchronous motor with P set as a pole; anintegrator 8 for calculating an AC phase θc in the control unit; a dqcoordinate converter 9 for converting a current value on a three-phaseAC axis into a component on a dc-qc axis as a rotation coordinate axis;an Iq* generator 10 for supplying a current command Id* of a d axiscomponent of the synchronous motor; an Iq* generator 11 for supplying acurrent command Iq* of a q axis component (torque component) of thesynchronous motor; a voltage command calculator 12 for calculatingvoltage commands Vdc* and Vqc* on a dc-qc axis based on ω1*, Id* andIq*; a dq reverse converter 13 for converting the voltage commands Vdc*and Vqc* on the dc-qc axis into values on the three-phase AC axis; anaxial error calculator 14 for estimating an axial error between a d-qaxis and a control axis dc-qc of the synchronous motor; a zero generator15 for supplying a zero command to the axial error; an adder 16 foradding or subtracting a signal; and a magnetic pole axis estimation gain17 for calculating an amount of correction to the electrical anglefrequency command ω1* by using the axial error. The inverter 4 includesa DC power source unit 41 constituting a main circuit power source ofthe inverter 4, a main circuit unit 42 of the inverter, a gate driver 43for generating a gate signal to the main circuit; a three-phase AC powersource 411 for supplying power to the inverter 4, a diode bridge 412 forrectifying the three-phase AC power source, and a smoothing capacitor413 for suppressing a pulsation component contained in a DC powersupply.

FIG. 2 is a perspective view showing a schematic structure of thesynchronous motor driving system of the first embodiment shown in FIG.1. The synchronous motor driving system of the present invention mainlyhas an AC power source unit, a control/inverter unit, and a synchronousmotor. As shown in FIG. 2, a control board includes the rotational speedcommand generator 1, the control unit 2, and the PWM generator 3, whichare all shown in FIG. 1. Actually, the control board is a digitalcircuit around a micro-processor. Also, the inverter main circuit 4, thecurrent detector 6 and the like are installed in one unit.

Next, description is made of an operation principle of the firstembodiment by referring to FIG. 1. The conversion gain 7 calculates anelectrical angle frequency ω1* of the synchronous motor based on arotational speed command ωr* from the rotational speed command generator1, and outputs it.

On the other hand, in the axial error calculator 14, an axial error Δθis estimated based on a current command and a detected current value.The magnetic pole axis estimation gain 17 calculates a speed correctionamount Δω1 based on the estimated axial error Δθ. The adder 16 adds theω1* and the Δω1 together to obtain ω1 c. The phase calculator 8integrates the ω1 c to obtain an AC phase θc in the control unit. As aresult, the AC phase θc is corrected by the Δθc. Then, bycoordinate-converting the detected value of the three-phase AC currentbased on this θc, an Idc as a dc axis component, and an Iqc as a qc axiscomponent are obtained. At the Id* generator 10 and the Iq* generator11, current commands of respective axis components of the synchronousmotor 5 are supplied. A method of generating the Id* and the Iq* will bedescribed in detail later.

The voltage command calculator 12 calculates voltages Vdc* and Vqc* tobe applied to the synchronous motor 5 based on the rotational speed ω1*and the current commands Id* and Iq* by an equation (2). In theequation, R: motor resistance, Ld: d axis inductance, Lq: q axisinductance, and Ke: generated power constant of motor.

V _(dc) *=R·I _(d)*−ω₁ L _(q) ·I _(q)*

V _(qc)*=ω₁ ·L _(d) ·I _(d) *+R·I _(q) *+K _(e)·ω₁  [Equation 2]

The equation (2) is similar to a calculation equation used for normalvector control. The equation (2) is described, for example as anequation (4.6) in a document 2: p 78, “Theory and Designing Practice ofAC Servo System”, by Hidehiko Sugimoto, Sogo Denshi Publishing (May,1990).

The dq reverse converter 13 coordinate-converts the voltages Vdc* andVqc* obtained by the equation (2) into voltage command values V1* on thethree-phase AC axis. Then, at the PWM generator 3, a voltage command V1*is converted into a pulse width. The gate driver 43 drives a switchingelement based on this pulse signal, and applies a voltage equal to eachof the voltages Vdc* and Vqc* to the synchronous motor 5.

FIG. 3 is a block diagram showing an internal configuration of thevoltage command calculator 12 in the synchronous motor driving system ofthe first embodiment of the present invention. The voltage commandcalculator 12 includes a gain 121 equivalent to a resistance value (R)of the synchronous motor, a gain 122 equivalent to a d axis inductance(Ld), a gain 123 equivalent to a q axis inductance (Lq), a multiplier124, and a gain 125 equivalent to a generated power constant (Ke).

As shown in the equation (2) and FIG. 3, a voltage command is calculatedby using constants R, Ld, Lq and Ke of the synchronous motor. If theseconstants of the synchronous motor are accurate, then the synchronousmotor is driven at a rotational speed indicated by a command value, anda current value.

By the axial error calculator 14, and the magnetic pole axis estimationgain 17, a phase locked loop (PLL) is formed and, by correcting the ω1*,a phase angle θc is corrected. As a result, an axial error Δθ iscontrolled to zero. Control response time for converging the axial errorto zero is decided by set response by the magnetic pole axis estimationgain 17. In addition, the magnetic pole axis estimation gain may be aproportional gain in the case of the configuration of the control systemof FIG. 1.

Next, description is made of an operation of the axial error calculator14 as a feature of the present invention. According to the document 1,an axial error Δθ can be calculated by an equation (3) using a voltagecommand, a detected current value, and a constant of the synchronousmotor.

[Equation 3]${\Delta\theta} = {\tan^{- 1}\frac{V_{d\quad c}^{*} - {R \cdot I_{d\quad c}} + {\omega_{1}{L_{q} \cdot I_{q\quad c}}}}{V_{q\quad c}^{*} - {R \cdot I_{q\quad c}} - {\omega_{1}{L_{q} \cdot I_{d\quad c}}}}}$

Codes Vdc* and Vqc* in the equation (3) represent parameters shown inthe equation (2), both of which strongly depend on an electrical anglefrequency ω1. If Id* and Iq* are constant, then Vdc* and Vqc* arechanged substantially in proportion to ω1*. Thus, when the ω1* is nearzero, a denominator/numerator of the equation (3) approaches zero,causing a considerable reduction in calculation accuracy.

When the ω1* is near zero, dependence of a term of resistance R isincreased. Since the resistance R is strongly affected by temperaturedependence, nonlinearity by a semiconductor device, or the like,accurate setting of the resistance R is difficult, making it impossibleto establish the equation (3). Consequently, it is very difficult toestimate an axial error in a wide speed range by using the equation (3).An equation (4) is obtained by substituting the equation (2) for theequation (3), and arranging it.

[Equation 4]${\Delta\theta} = {\tan^{- 1}\frac{{\omega_{1}{L_{q}\left( {I_{q\quad c} - I_{q}^{*}} \right)}} - {R\left( {I_{d\quad c} - I_{d}^{*}} \right)}}{{K_{e} \cdot \omega_{1}} - {\omega_{1}\left( {{L_{q}I_{d\quad c}} - {L_{d}I_{d}^{*}}} \right)} - {R\left( {I_{q\quad c} - I_{q}^{*}} \right)}}}$

In a denominator/numerator of the equation (4), it can be seen thatthere are two types of terms, i.e., for ω1 and R.

An R term of the numerator (=−R(Idc−Id*)) indicates an amount of voltagereduced according to deviation between Idc and Id*. The Id is normallycontrolled to zero in the synchronous motor of a non-salient type. Also,in the case of that of a salient type, in a stationary state, changesonly occur in a range of 20 to 30% of synchronous motor rating. Thus,the amount of reduced voltage of this term becomes a small value ofabout 1% or lower, which can be ignored.

An R term of the denominator (=−R(Iqc−Iq*) indicates a voltage reductionof 1% or lower even in rated current, which can be ignored substantiallyin all speed range compared with a term of Ke·ω1. Moreover, as higherefficiency is strongly demanded for the synchronous motor in recentyears, the resistance R of the synchronous motor tends to be designedsmaller and smaller.

For the above-described reasons, by ignoring the R term in the equation(4), the equation (4) can be simplified to be an equation (5). That is,assuming that a d axis inductance is Ld[H], a q axis inductance Lq[H], agenerated power constant of the motor Ke[Wb], a current command value ofa magnetic pole d axis Id*, a current command on a q axis orthogonal tothe magnetic pole d axis Iq*, a detected current value on the assumed dcaxis of the magnetic pole axis Idc, and a detected current value on theqc axis orthogonal to the dc axis Iqc, an axial error Δθ is calculatedby the equation (5) using the current commands Id* and Iq* on the d-qaxis, and the detected current values Idc, Iqc on the dc-qc axis.

[Equation 5]${\Delta\theta} = {\tan^{- 1}\frac{L_{q}\left( {I_{q\quad c} - I_{q}^{*}} \right)}{K_{e} - \left( {{L_{q}I_{d\quad c}} - {L_{d}I_{d}^{*}}} \right)}}$

In the equation (5), the ω1 term is cancelled, no electrical anglefrequencies are present, and an axial error Δθ can be calculated. As theresistance of the equation (4) is ignored, an estimation error isgenerated in a very low speed zone of 1 to 2% or lower. However, it canbe detected whether there is an axial error. Even if there is a smallestimation error, since an axial error Δθ can be reduced to zero at theend, a vector control sensorless system can be achieved in a range ofsubstantially all speeds.

In the very low speed zone of 1 to 2% or lower, estimation of an axialerror is inevitably difficult. Thus, for example, it is impossible tooutput rated torque at a zero speed. However, passage can be allowedthrough the very low speed zone during acceleration/deceleration of thesynchronous motor.

FIG. 4 is a block diagram showing an internal configuration of the axialerror calculator 14 in the synchronous motor driving system of the firstembodiment of the present invention. That is, FIG. 4 shows theconfiguration of the axial error calculator 14 using the equation (5).The axial error calculator 14 includes a generated power constant setter126, and an arc tangent calculator 127.

The generated power constant setter 126 outputs a generated powerconstant (Ke), and the arc tangent calculator 127 calculates an arctangent of Y0/X0 for two inputs X0 and Y0. As shown in FIG. 4, byentering current commands Id* and Iq*, and detected current values Idcand Iqc, it is possible to calculate an axial error Δθ without anydependence on ω1.

In the equation (5), by using an approximation of tan(x)=x in a range ofsmall x, an axial error can be calculated in approximation by anequation (6).

[Equation 6]${\Delta\theta} = \frac{L_{q}\left( {I_{q\quad c} - I_{q}^{*}} \right)}{K_{e} - \left( {{L_{q}I_{d\quad c}} - {L_{d}I_{d}^{*}}} \right)}$

In addition, in the case of the synchronous motor of the non-salienttype, in the equation (5) or (6), by replacing Ld and Lq by oneinductance, an axial error can be calculated more easily. In the case ofthe synchronous motor of the non-salient type, since Id is normallycontrolled to be zero, calculation can be simplified more by setting Id*to Id*=0 in each calculation equation.

Next, more detailed description is made of the equations (5) and (6) tosimplify the axial error calculation equation more. Terms of numeratorsin both equations relate to a torque current Iq, and are greatly changedby load fluctuation or the like of the synchronous motor. In otherwords, it can be understood that the numerator terms greatly affectpresence of axial errors and error directions (polarities).

On the other hand, terms of denominators relate to a generated powerconstant Ke and Id. A code Ke denotes a magnetic flux of a permanentmagnet, which is larger than that generated by Id. Thus, Ke is dominantnormally. Therefore, even if Id is slightly changed during transition,great fluctuation of a denominator value may be limited. Therefore, whenpriority is placed on simplicity of calculation rather than on preciseaxial error calculation, Id* can be used instead of Idc. In such a case,the equation (5) is changed to an equation (7).

[Equation 7]${\Delta\theta} = {\tan^{- 1}\frac{L_{q}\left( {I_{q\quad c} - I_{q}^{*}} \right)}{K_{e} - {\left( {L_{q} - L_{d}} \right)I_{d}^{*}}}}$

By using the equation (7), an axial error can be calculated more easily,making it possible to reduce a load on control calculation.

Second Embodiment

FIG. 5 is a block diagram showing a configuration of a synchronous motordriving system according to a second embodiment of the presentinvention. The synchronous motor driving system of the second embodimentincludes an inverter 4B, a DC current detector 44 for detecting acurrent I0 flowing from a DC power supply unit 41 to an inverter maincircuit unit, and a synchronous motor current estimator 18 forestimating a synchronous motor current based on a detected value of thecurrent I0.

In the second embodiment of FIG. 5, the current detector 6 is removedfrom the configuration of the system shown in FIG. 1, and instead the DCcurrent detector 44, and the synchronous motor current estimator 18 areadded. The DC current detector 44 detects a current by using HALL CT orshunt resistance. The synchronous motor current estimator 18 estimates asynchronous motor current based on the detected value of the current I0,and a pulse waveform outputted from a PWM generator 3.

A control unit 2 controls the synchronous motor by regarding anestimated value I1 c as a detected current value of the synchronousmotor. Thus, an operation of the control unit 2 itself is similar tothat of the first embodiment.

FIG. 6 is explanatory view showing an operation of the synchronous motorcurrent estimator 18 in the synchronous motor driving system of thesecond embodiment of the present invention. In FIG. 6, each of (a) to(c) shows a PWM pulse waveform of each phase. At 1, a switch (Sup, Svpor Swp) of a plus side is turned ON and, at 0, a switch (Sun, Svn orSwn) of a minus side is turned ON.

Now, assuming that a synchronous motor current is similar to that shownin FIG. 6(d), the DC current I0 of the inverter has a waveform similarto that shown in FIG. 6(e). For the waveform of FIG. 6(e), there arefour modes described below.

(1) Mode 1:

Sup=ON, Svp=ON, Swp=ON→I0=0

(2) Mode 2:

Sup=ON, Svp=ON, Swp=OFF→I0=Iu+Iv=−Iw

(3) Mode 3:

Sup=ON, Svp=OFF, Swp=OFF→I0=Iu

(4) Mode 4:

Sup=OFF, Svp=OFF, Swp=OFF→I0=0

Thus, Iu can be detected if the DC current I0 is detected in theswitching state of the mode 3 and, in the state of the mode 2, Iw can bedetected. Iv may be calculated based on Iu and Iw. A basic operation ofthe synchronous motor current estimator is similar to that of a methoddisclosed in, for example JP-A-6-153526 or JP-A-8-19263.

However, there was a big problem when this synchronous motor currentestimator was used for the conventional vector control sensorlesssystem. As described above, in order to achieve the vector controlsensorless system in a low-speed zone, the method using the saliency ofthe synchronous motor, i.e., the method of estimating an axial errorfrom a higher harmonic wave component of a current by superposing ahigher harmonic wave on a voltage, is only available. Thus, it wasnecessary to accurately detect a higher harmonic wave flowing to thesynchronous motor.

In the configuration of the control system of FIG. 5, as shown in FIG.6, because of dependence of a timing for current detection on a state ofthe PWM pulse, it is difficult to accurately detect a higher harmonicwave component in a current. A basic wave component can be detectedwithout any problems, since a frequency is sufficiently lower comparedwith a PWM pulse frequency. Accordingly, detection of the higherharmonic wave is a problem. To increase accuracy of detection,improvements such as an increase in the amount of superposing a higherharmonic wave, or a reduction in a frequency of the superposed wave.Either case may reduce efficiency, causing a great increase in noise.

On the other hand, in the axial error calculation of the presentinvention, it is not necessary to superpose any higher harmonic wavesand, by using the equation (5) or (6), accurate axial error calculationis possible in substantially all speed zones. Therefore, ahigh-performance can be expanded to substantially all the zones.

Third Embodiment

The above-described second embodiment enables the number of currentsensors to be reduced, and thus provides an advantage of simplifying theconfiguration of the control system. However, the following problems areinherent. That is, when a rotational speed of the synchronous motor islow, and an output voltage is small, the periods of the modes 2 and 3shown in FIG. 6 become short, necessitating reading of a very narrowpulse-like current. Waveforms in FIG. 6 are for principle explanation,and I0 represents a staircase having no vibration. In practice, however,ringing is superposed in a current waveform following switching. If apulse width is narrow, this effect cannot be ignored.

FIG. 7 is a block diagram showing a configuration of a synchronous motordriving system according to a third embodiment of the present invention,which is provided to solve the problems of the second embodiment. Thesynchronous motor driving system of the third embodiment includes acontrol unit 2C, a filter 19 for removing a pulsation componentcontained in a DC current I0, and an Iqc estimator 20 for estimating a qaxis current Iqc on a control axis.

The third embodiment of FIG. 7 is substantially similar in systemconfiguration to the second embodiment of FIG. 5. However, thesynchronous motor current estimator 18 of FIG. 5 is removed and,instead, the filter 19 and the Iqc estimator 20 are added.

In the control unit 2C, compared with the first embodiment of FIG. 1,the dq coordinate converter 9 is removed, and a system configuration isemployed, where Idc and Iqc are not calculated from a synchronous motorcurrent.

Instead, Iqc necessary for control is obtained by using the Iqcestimator 20. No detection/estimation of Idc is carried out.Accordingly, an axial error calculator 14 estimate an axial error Δθaccording to the equation (7) without using Idc.

Next, description is made of an operation principle for the filter 19and the Iqc estimator 20. The filter 19 removes a PWM pulse componentfrom a DC current I0, and extracts an average value of I0. This filteris provided for the purpose of removing a carrier frequency component.Accordingly, a cutoff frequency of the filter only needs to be set equalto about 1 of several, or 1 of several tens of a carrier frequency.Thus, an effect of ringing following switching can be completelyremoved. As a result, from an output of the filter 19, a DC current I0having a higher harmonic wave removed is obtained. The Iqc estimator 20estimates Iqc by using the current I0 having the higher harmonic waveremoved.

Next, description is made of a principle of the Iqc estimation. Arelation between a voltage/current on a d-q axis of the synchronousmotor, and a DC power supply voltage V0 and the DC current I0 of aninverter is represented by an equation (8) with regard to power.

[Equation 8]${I_{0}V_{0}} = {\frac{3}{2}\left( {{V_{d}I_{d}} + {V_{q}I_{q}}} \right)}$

A coefficient 3/2 in the right side represents a coefficient whenrelative conversion is used as d-q coordinate conversion. In the case ofabsolute conversion, the coefficient becomes 1. Since the right side ofthe equation (8) is established in any coordinate, a relation on a dc-qcaxis can be represented by an equation (9).

[Equation 9]${I_{0}V_{0}} = {\frac{3}{2}\left( {{V_{d\quad c}I_{d\quad c}} + {V_{q\quad c}I_{q\quad c}}} \right)}$

Assuming that the inverter is ideal, Vdc and Vqc can be replaced by Vdc*and Vqc*, and voltage commands can be used instead. Iqc obtained fromthe equation (9) is represented by an equation (10).

[Equation 10]$I_{q\quad c} = \frac{{\frac{2}{3}I_{0}V_{0}} - {V_{d\quad c}^{*}I_{d\quad c}}}{V_{q\quad c}^{*}}$

In the equation (10), since Idc cannot be detected, if a command valueId* is used instead, a relation is represented by an equation (11).

[Equation 11]$I_{q\quad c} = \frac{{\frac{2}{3}I_{0}V_{0}} - {V_{d\quad c}^{*}I_{d}^{*}}}{V_{q\quad c}^{*}}$

When the Idc is replaced by Id*, an estimation error may be slightlyincreased. However, since a q axis (qc axis) is dominant in an output ofthe synchronous motor, no large errors are generated. The Iqc estimatorestimates Iqc by using calculation of the equation (11). A DC voltage V0may be directly detected by using a sensor. However, if fluctuation in aDC voltage is small, a set value (command value) of the DC current canbe used.

In addition, since the equation (8) is a relational equation whenconversion efficiency of the inverter is assumed to be 1, the estimatedvalue includes an error due to the assumption. Thus, to increaseaccuracy of estimation, Iqc may be estimated by considering theconversion efficiency of the inverter.

According to the third embodiment, without using any current sensors ofthe synchronous motor, it is possible to achieve a synchronous motordriving system of a vector control sensorless system, which is simplerin configuration, and higher in performance.

Fourth Embodiment

FIG. 8 is a block diagram showing an internal configuration of an axialerror calculator 14D in a synchronous motor driving system according toa fourth embodiment of the present invention. In the fourth embodiment,instead of the axial error calculator 14 in the first or thirdembodiment, the axial error calculator 14D including a functiongenerator 21 for generating a function shown in FIG. 9 is used.

Axial error calculation of the present invention can be achieved byremoving the R term of the equation (4) as described above. In thismethod of calculation, when a rotational speed was extremely low, i.e.,1 to 2% or lower, there was a possibility of an error generated in aresult of axial error calculation. Thus, the equation of calculation iscorrected in order to increase accuracy of axial error calculation in avery low speed zone. By modifying the equation (4), a relation isrepresented by an equation (12).

[Equation 12]${\Delta\theta} = {\tan^{- 1}\frac{{L_{q}\left( {I_{q\quad c} - I_{q}^{*}} \right)} - \frac{R\left( {I_{d\quad c} - I_{d}^{*}} \right)}{\omega_{1}}}{K_{e} - \left( {{L_{q}I_{d\quad c}} - {L_{d}I_{d}^{*}}} \right) - \frac{R\left( {I_{q\quad c} - I_{q}^{*}} \right)}{\omega_{1}}}}$

In the equation (12), it can be understood that as ω1 is smaller, aneffect of an R term is larger.

However, direct use of the equation (12) may cause a considerableincrease in a calculation error when ω1 is very small. In a worst case,division by zero may even occur. Thus, the equation (12) is modified asfollows:

[Equation 13]${\Delta\theta} = {\tan^{- 1}\frac{{L_{q}\left( {I_{q\quad c} - I_{q}^{*}} \right)} - {{K_{r\quad x}\left( \omega_{1} \right)}\left( {I_{d\quad c} - I_{d}^{*}} \right)}}{K_{e} - \left( {{L_{q}I_{d\quad c}} - {L_{d}I_{d}^{*}}} \right)}}$

Since a term of Ke is dominant in denominator term of the equation (12),correction is made only for a numerator by a function Krx.

FIG. 9 illustrates an operation of a function generator in thesynchronous motor driving system of the fourth embodiment of the presentinvention. A function Krx is set, for example, similar to that shown inFIG. 9. In a region where ω1* is equal to/higher than ω1L, Krx=0 is set,and a term for R is removed. Only when ω1* is equal to/lower than ω1L,axial error calculation is corrected to increase accuracy of axial errorestimation. However, a function Krx at time of ω1*=0 is limited to afinite value (Krx0), and problems of calculation such as division byzero are prevented. As a result, even in the very low speed zone, it ispossible to increase the accuracy of axial error calculation, andachieve a synchronous motor driving system of a vector controlsensorless system in a wider range.

Fifth Embodiment

FIG. 10 is a block diagram showing an internal configuration of acontrol unit 2E in an synchronous motor driving system according to afifth embodiment of the present invention. In the fifth embodiment,instead of the Iq* generator 11 for supplying a current command Iq* of aq axis component (torque component) of the synchronous motor, an Iq*generator 11E is provided. The Iq* generator 11E of the fifth embodimentcalculates a q axis current command Iq* based on a detected currentvalue Iqc.

In the case of vector control, it is always necessary to control avoltage applied to the synchronous motor and a current of thesynchronous motor to a relation represented by the equation (2). Id* hasno direct relation to a load on the synchronous motor, and thus can beset to an optional value. On the other hand, Iq* must be properlychanged according to load torque, and a rotational speed.

In a stationary state, a relation of Iq*=Iqc must always be set.Otherwise, an axial error Δθ is left, making it impossible to establishvector control. According to the fifth embodiment of the presentinvention, by an extremely simple system configuration, Iq* can bematched with Iqc. The Iq* generator 11E calculates an equation (14).Here, a code Tr denotes a time constant; and s Laplacian operator.

[Equation 14]$I_{q}^{*} = {\frac{1}{1 + {T_{r} \cdot s}} \cdot I_{q\quad c}}$

The equation (14) represents a first order lag element and, inprinciple, a stationary value of Iqc is set as Iq*. Accordingly, Iqc=Iq*is established at the end, establishing vector control.

A current command in normal control is supplied before a real detectedcurrent value, and the detected value is matched with the currentcommand. However, in the system configuration of FIG. 10, different fromthe normal case, a command is later matched with a necessary currentvalue, i.e., an actually flowing current value, and thereby balance iskept between a voltage and a current.

By using the control unit of FIG. 10, there are two places where controlconstants are adjusted, i.e., a magnetic pole axis estimation gain 17,and the time constant of the equation (14). The system configuration isthus simplified greatly, realizing vector control. Moreover, since theaxial error calculator 14 enables high-performance control to beachieved in range of substantially all speeds, there is problem ofsystem switching for each speed zone, and it is possible to achieve asynchronous motor driving system of a vector control sensorless systemhaving the number of places to be adjusted set to a minimum.

FIG. 11 is a block diagram showing an internal configuration of acontrol unit 2F when the synchronous motor driving system of the fifthembodiment of the present invention is applied to the third embodimentof FIG. 7. In FIG. 11, an Iq* generator 11E fetches an estimateddetected current value not from the dq coordinate converter 9 of FIG. 10but from the Iqc estimator 20 of FIG. 7. Also in this case, a q axiscurrent command Iq* is calculated based on a detected current value Iqc.Thus, by a simpler system configuration, it is possible to achieve asynchronous motor driving system of a vector control sensorless system.

Sixth Embodiment

FIG. 12 is a block diagram showing an internal configuration of acontrol unit 2G in a synchronous motor driving system according to asixth embodiment of the present invention. The control unit 2G of thesixth embodiment includes an Id current control unit 22 for controllinga d axis current, an Iq current control unit 23 for controlling a q axiscurrent, a code inverter 24 for inverting a code of Δω1, a conversiongain 25 for converting Δω1 into speed deviation with P set as a pole ofthe synchronous motor, and a speed control unit 26 for setting speeddeviation to zero.

As described above with reference to the fifth embodiment, in the vectorcontrol, making of a current command Iq* is extremely important. In thefifth embodiment, since a current command Iq* is obtained from a realdetected current value (estimated value), it is very easy. However,high-speed response to speed or load fluctuation is difficult.

On the other hand, by using the control unit 2G shown in FIG. 12, it ispossible to achieve a synchronous motor system of a vector controlsensorless system capable of making high-speed response.

An output Δω1 of a magnetic pole axis estimation gain 17 is a correctedspeed amount for reducing an axial error to zero. In other words, thisoutput is an amount corresponding to deviation between a real rotationalspeed command ωr* and a real speed ωr. Thus, by supplying Iq* so as toreduce the output Δω1 to zero, response of the speed control of thesynchronous motor can be improved.

The output Δω1 has its polarity inverted by the code inverter 24,multiplied by 2/P by the conversion gain 25, and speed deviation Δωr(=ωr*−ωr) is obtained. The speed control unit 26 includes aproportional/integration compensation element, and the like, andcalculates a torque current command Iq* based on the speed deviationΔωr.

Moreover, to improve a response characteristic of the synchronous motor,current control units 22 and 23 are added to dc, and qc axes, and acurrent is controlled at a high speed.

As a result, it is possible to achieve a synchronous motor drivingsystem capable of making high-speed response to speed fluctuation,external torque disturbances, and the like. In addition, since an axialerror calculator can be applied in a range of substantially all speeds,it is possible to achieve a synchronous motor driving system of a vectorcontrol sensorless system having control performance considerablyimproved compared with the conventional vector control sensorlesssystem.

Seventh Embodiment

FIG. 13 is a block diagram showing an internal configuration of acontrol unit 2H in a synchronous motor driving system according to aseventh embodiment of the present invention. In FIG. 13, a magnetic poleaxis estimation gain 17H includes a proportion/integration compensationelement. The control unit 2H of FIG. 13 is substantially similar insystem configuration to the control unit 2G of the sixth embodiment, butdifferent in a method of making ω1 in the control unit 2H.

In the foregoing first to sixth embodiments, ω1* was directly calculatedfrom the rotational speed command ωr*, Δω1 was added, and the drivingfrequency was corrected.

On the other hand, the seventh embodiment has a feature that an outputof a magnetic pole axis estimation gain is set to be ω1, used forcontrol calculation.

A speed control unit 26 calculates Iq* based on deviation Δω between arotational speed command ωr* and a real speed (estimated value) ωr. TheIq* is compared with a real current value Iqc, and a current iscontrolled by an Iq current control unit 23 such that both can bematched with each other. When a torque current is actually generated inthe synchronous motor, and a rotational speed of the synchronous motoris changed to generate an axial error, an axial error calculator 14detects this axial error. The magnetic pole axis estimation gain 17Hreceives the axial error, corrects the ω1 and outputs it.

With the control system constructed in such a manner, it is possible toperform acceleration/deceleration of the synchronous motor by maximumtorque. A current to be supplied to the synchronous motor is limited bysynchronous motor rating or an inverter capacity. By performingacceleration/deceleration by a maximum condition in this range, it ispossible to control the synchronous motor at a highest speed.

Accordingly, a limiter of a maximum current is provided in the Iq* and,in a maximum flowing state of a torque current, the synchronous motor isaccelerated. In this case, a speed ω1 must be a speed resulted fromapplication of torque and, different from the case shown in FIG. 12, aspeed cannot be provided from a rotational speed command in a feedforward manner. Thus, when the synchronous motor isaccelerated/decelerated by maximum torque, the system configuration ofFIG. 13 is necessary.

By employing the axial error calculator 14 of the seventh embodiment, itis possible to achieve a high-speed acceleration/decelerationcharacteristic in a wide range.

The control units 2G and 2H of FIGS. 12 and 13 can be applied to thesystem of a configuration shown FIG. 5 or 7. If applied to the systemconfiguration of FIG. 5, the control unit only needs to be directlyreplaced. If applied to the system configuration of FIG. 7, the filter19 and the Iqc estimator 20 need to be provided in the control unit.However, since Idc cannot be detected in the system configuration ofFIG. 7, no Id current control units are installed.

According to the seventh embodiment, it is possible to achieve asynchronous motor driving system having control performance considerablyimproved compared with a high-performance sensorless system havingfurther improved high-speed responsiveness.

Eighth Embodiment

FIG. 14 is a block diagram showing an internal configuration of acontrol unit 2J in a synchronous motor driving system according to aneighth embodiment of the present invention. In the eighth embodiment,instead of the Id* generator 10 in the first to seventh embodiments, anId* generator 10J is used. The Id* generator 10J of FIG. 14 decides avalue of Id* based on Iq*. That is, the eighth embodiment issubstantially similar to the control unit of the fifth embodiment shownin FIG. 10, but has a feature in a method of making a current commandId*.

Among permanent magnet synchronous motors, there is a type, whichgenerates synchronous motor torque by combining torque generated by apermanent magnet with reluctance torque generated by saliency (reversesaliency) of the synchronous motor. In the case of the synchronous motorof this type, a maximum torque point of the synchronous motor is locatedin a region where Id is controlled to be a minus value, and control ofId=0 is not advantageous for efficiency. Thus, to drive the synchronousmotor by maximum efficiency, the synchronous motor is preferably drivenalways by maximum torque. Especially, in an industrial/home electricappliance field, energy conservation has been demanded, and maximizationof efficiency is an important task.

Conditions for obtaining maximum torque are described in, for example ina document 3: pp. 662-667, “Comparison of Control Characteristics ofPermanent Magnet Synchronous Motors with Several Rotor Configurations”,JIEE papers D, Vol. 114-6, 1994, or the like. According to an equation(6) of the document 3, a relation is represented by an equation (15).When Iq is set, Id for obtaining maximum torque is decided. However, Φmdenotes a magnetic flux of a permanent magnet, and Ld≠Lq is established.

[Equation 15]$I_{d} = {\frac{\Phi_{m}}{2\left( {L_{q} - L_{d}} \right)} - \sqrt{\frac{\Phi_{m}^{2}}{4\left( {L_{q} - L_{d}} \right)^{2}} + I_{q}^{2}}}$

In the eighth embodiment, the Id* generator calculates the equation (15)by using Id*. As a result, it is always possible to drive thesynchronous motor by maximum torque (maximum efficiency). For thecalculation of the equation (15), Iqc may be used instead of Id*.However, since fluctuation is large in the Iqc during transition, theentire control system may become unstable.

Efficiency maximization can contribute to energy conservation of theapparatus if functioning in s stationary state, and thus use of Iq* asan output of the Iq* generator causes no problems.

Moreover, application of the Id* generator 10J of the eighth embodimentto the other first to seventh embodiments causes no problems.

Thus, by using the eighth embodiment, it is possible to provide asynchronous motor system of a vector control sensorless system, which iscapable of operating the synchronous motor by maximum efficiency.

According to the present invention, it is possible to provide thesynchronous motor driving system capable of achieving the vector controlsensorless system to cover the wide range from the low to high speedzone without any reductions in efficiency or increases in noise.Moreover, irrespective of presence of saliency of the synchronous motorto be controlled, it is possible to achieve the synchronous motordriving system of the high-performance, and highly accurate vectorcontrol sensorless system.

It should be further understood by those skilled in the art that theforegoing description has been made on embodiments of the invention andthat various changes and modifications may be made in the inventionwithout departing from the spirit of the invention and the scope of theappended claims.

What is claimed is:
 1. A synchronous motor driving system whichcomprises a synchronous motor, an inverter for driving the synchronousmotor, a rotational speed command generator for supplying a rotationalspeed command to the synchronous motor, and a control unit forcalculating a voltage applied to the synchronous motor, said synchronousmotor driving system comprising axial error calculation means forestimating an axial error Δθ between a d-q axis and a dc-qc axis byusing Ld, Lq, Ke, Id*, Iq*, Idc and Iqc in a range of all rotationalspeeds except zero of the rotational speed command of the synchronousmotor wherein Ld is an inductance on a magnetic pole axis d, Lq is aninductance on a q axis orthogonal to the magnetic pole axis d, Ke is agenerated power constant of the motor, Id* is a current command of the daxis, Iq* is a current command on the q axis, Idc is a detected currentvalue on an assumed dc axis on control, and Iqc is a detected currentvalue on an assumed qc axis orthogonal to the assumed dc axis; and meansfor adjusting the dc-qc axis to the d-q axis based on the calculatedvalue of the axial error Δθ.
 2. The synchronous motor driving systemaccording to claim 1, wherein the axial error calculation means is meansfor calculating an axial error Δθ in accordance with an equation (1) byusing current commands Id* and Iq* on the d-q axis, and detected currentvalues Idc and Iqc on the dc-qc axis, wherein Ld is an inductance on amagnetic pole axis d, Lq is an inductance on a q axis orthogonal to themagnetic pole axis d, Ke is a generated power constant of the motor, Id*is a current command of the d axis, Iq* is a current command on the qaxis, Idc is a detected current value on an assumed dc axis on control,and Iqc is a detected current value on an assumed qc axis orthogonal tothe assumed dc axis: [Equation 1]${\Delta\theta} = {\tan^{- 1}{\frac{L_{q}\left( {I_{q\quad c} - I_{q}^{*}} \right)}{K_{e} - \left( {{L_{q}I_{d\quad c}} - {L_{d}I_{d}^{*}}} \right)}.}}$


3. The synchronous motor driving system according to claim 1, furthercomprising means for detecting a DC current on a power source side ofthe inverter, and synchronous motor current estimating means forestimating an AC current of the synchronous motor based on the detectedDC current and a driving pulse signal for driving the inverter, theaxial error Δθ being calculated using the estimated current as adetected current value.
 4. The synchronous motor driving systemaccording to claim 1, further comprising means for detecting a DCcurrent on a power source side of the inverter, and Iqc estimating meansfor estimating a current value on the qc axis of the synchronous motorbased on the detected DC current, and a detected value or a set value ofa DC voltage of the inverter, the axial error Δθ being calculated usingthe estimated current as a detected current value.
 5. The synchronousmotor driving system according to claim 1, further comprising acorrection term for making correction in accordance with a rotationalspeed command of the synchronous motor in the calculation of the axialerror Δθ, said correction term being a function of weight whichincreases as the rotational speed command approaches zero.
 6. Thesynchronous motor driving system according to claim 1, wherein thecurrent command Iq* on the q axis is made based on the detected currentvalue or the estimated value on the qc axis.
 7. The synchronous motordriving system according to claim 1, wherein the current command Iq* onthe q axis is made based on the detected current value or the estimatedvalue on the qc axis.
 8. The synchronous motor driving system accordingto claim 1, further comprising means for estimating a speed deviationbetween the rotational speed command and a real rotational speed basedon the calculated value of the axial error Δθ, the q axis currentcommand Iq* of the synchronous motor being made based on the estimatedvalue of the speed deviation.
 9. The synchronous motor driving systemaccording to claim 6, wherein the current command Id* of the d axis ismade based on the current command Iq* of the q axis.
 10. The synchronousmotor driving system according to claim 7, wherein the current commandId* of the d axis is made based on the current command Iq* of the qaxis.
 11. The synchronous motor driving system according to claim 8,wherein the current command Id* of the d axis is made based on thecurrent command Iq* of the q axis.
 12. The synchronous motor drivingsystem according to claim 1, wherein said synchronous motor is of anon-salient type.
 13. A synchronous motor driving system which comprisesa synchronous motor, an inverter for driving the synchronous motor, arotational speed command generator for supplying a rotational speedcommand to the synchronous motor, and a control unit for calculating avoltage applied to the synchronous motor, said synchronous motor drivingsystem comprising: axial error calculator for estimating an axial errorΔθ between a d-q axis and a dc-qc axis by using Ld, Lq, Ke, Id*, Iq*,Idc and Iqc in a range of all rotational speeds except zero of therotational speed command of the synchronous motor wherein Ld is aninductance on a magnetic pole axis d, Lq is an inductance on a q axisorthogonal to the magnetic pole axis d, Ke is a generated power constantof the motor, Id* is a current command of the d axis, Iq* is a currentcommand on the q axis, Idc is a detected current value on an assumed dcaxis on control, and Iqc is a detected current value on an assumed qcaxis orthogonal to the assumed dc axis; and a circuit for adjusting thedc-qc axis to the d-q axis based on the calculated value of the axialerror Δθ.
 14. The synchronous motor driving system according to claim13, further comprising a detector for detecting a DC current on a powersource side of the inverter, and a synchronous motor current estimatorfor estimating an AC current of the synchronous motor based on thedetected DC current and a driving pulse signal for driving the inverter,the axial error Δθ being calculated using the estimated current as adetected current value.
 15. The synchronous motor driving systemaccording to claim 13, further comprising: a detector for detecting a DCcurrent on a power source side of the inverter, and an Iqc estimator forestimating a current value on the qc axis of the synchronous motor basedon the detected DC current, and a detected value or a set value of a DCvoltage of the inverter, the axial error Δθ being calculated using theestimated current as a detected current value.
 16. The synchronous motordriving system according to claim 13, further comprising a correctionterm for making correction in accordance with a rotational speed commandof the synchronous motor in the calculation of the axial error Δθ, saidcorrection term being a function of weight which increases as therotational speed command approaches zero.
 17. The synchronous motordriving system, according to claim 13, wherein the current command Iq*on the q axis is made based on the detected current value or theestimated value on the qc axis.
 18. The synchronous motor driving systemaccording to claim 13, further comprising an estimator for estimating aspeed deviation between the rotational speed command and a realrotational speed based on the calculated value of the axial error Δθ,the q axis current command Iq* of the synchronous motor being made basedon the estimated value of the speed deviation.
 19. The synchronous motordriving system according to claim 13, further comprising an estimatorfor estimating a rotational speed of the synchronous motor based on thecalculated value of the axial error Δθ, the q axis current command Iq*of the synchronous motor being made based on a deviation between theestimated value and the rotational speed command.
 20. The synchronousmotor driving system according to claim 17, wherein the current commandId* of the d axis is made based on the current command Iq* of the qaxis.
 21. The synchronous motor driving system according to claim 18,wherein the current command Id* of the d axis is made based on thecurrent command Iq* of the q axis.
 22. The synchronous motor drivingsystem according to claim 19, wherein the current command Id* of the daxis is made based on the current command Iq* of the q axis.
 23. Thesynchronous motor driving system according to claim 13, wherein saidsynchronous motor is of a non-salient type.
 24. The synchronous motordriving system according to claim 13, wherein the axial errorcalculation means is means for calculating an axial error Δθ inaccordance with an equation (1) by using current commands Id* and Iq* onthe d-q axis, and detected current values Idc and Iqc on the dc-qc axis,wherein Ld is an inductance on a magnetic pole axis d, Lq is aninductance on a q axis orthogonal to the magnetic pole axis d, Ke is agenerated power constant of the motor, Id* is a current command of the daxis, Iq* is a current command on the q axis, Idc is a detected currentvalue on an assumed dc axis on control, and Iqc is a detected currentvalue on an assumed qc axis orthogonal to the assumed dc axis: [Equation1]${\Delta\theta} = {\tan^{- 1}{\frac{L_{q}\left( {I_{q\quad c} - I_{q}^{*}} \right)}{K_{e} - \left( {{L_{q}I_{d\quad c}} - {L_{d}I_{d}^{*}}} \right)}.}}$


25. The synchronous motor driving system according to claim 13, whereinthe axial error calculation means is means for calculating an axialerror Δθ in accordance with an equation (1) by using current commandsId* and Iq* on the d-q axis, and detected current value Iqc on the dc-qcaxis, wherein Ld is an inductance on a magnetic pole axis d, Lq is aninductance on a q axis orthogonal to the magnetic pole axis d, Ke is agenerated power constant of the motor, Id* is a current command of the daxis, Iq* is a current command on the q axis, and Iqc is a detectedcurrent value on an assumed qc axis orthogonal to an assumed dc axis oncontrol: [Equation 1]${\Delta\theta} = {\tan^{- 1}{\frac{L_{q}\left( {I_{q\quad c} - I_{q}^{*}} \right)}{K_{e} - \left( {{L_{q}I_{d\quad c}} - {L_{d}I_{d}^{*}}} \right)}.}}$


26. The synchronous motor driving system according to claim 1, whereinthe axial error calculation means is means for calculating an axialerror Δθ in accordance with an equation (1) by using current commandsId* and Iq* on the d-q axis, and detected current value Iqc on the dc-qcaxis, wherein Ld is an inductance on a magnetic pole axis d, Lq is aninductance on a q axis orthogonal to the magnetic pole axis d, Ke is agenerated power constant of the motor, Id* is a current command of the daxis, Iq* is a current command on the q axis, and Iqc is a detectedcurrent value on an assumed qc axis orthogonal to an assumed dc axis oncontrol: [Equation 1]${\Delta\theta} = {\tan^{- 1}{\frac{L_{q}\left( {I_{q\quad c} - I_{q}^{*}} \right)}{K_{e} - \left( {{L_{q}I_{d\quad c}} - {L_{d}I_{d}^{*}}} \right)}.}}$